Hysteretic control of a boost converter

ABSTRACT

A boost converter includes a clock generator, a switching converter, a hysteretic controller, and a power tracking module. The clock generator configured to output a clock signal; The switching converter configured to operate at a frequency based on the clock signal. The hysteretic controller configured to regulate an intermediate output from the switching converter. The power tracking module configured to change a frequency control signal that is sent to the clock generator, the change in frequency is based on a current flowing into an output capacitor such that a charge time of the capacitor is minimized when the current is maximized.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. provisional application Ser.No. 62/889,509 filed Aug. 20, 2019, the disclosure of which is herebyincorporated in its entirety by reference herein.

TECHNICAL FIELD

This invention relates generally to a system and method of hystereticcontrol feedback to operate a boost converter.

BACKGROUND

An energy harvester has a transducer that converts physical energy likevibration, light or temperature difference to electrical energy. Thiselectrical energy is often generated at a voltage that is different fromthe voltage required to charge a battery or a capacitor. A switchingvoltage converter is generally used to step up/down the voltage.

Energy harvesting systems generally regulate output voltage withhysteretic control due to its simplicity and low power implementation.However, tracking maximum output power under varying inputs ischallenging in energy-constrained applications. This disclosuredescribes an improved modified hysteretic controller with higherefficiency that also provides the control signals for tracking outputpower.

SUMMARY

A boost converter includes a clock generator, a switching converter, ahysteretic controller, and a power tracking module. The clock generatorconfigured to output a clock signal; The switching converter configuredto operate at a frequency based on the clock signal. The hystereticcontroller configured to regulate an intermediate output from theswitching converter. The power tracking module configured to change afrequency control signal that is sent to the clock generator, the changein frequency is based on a current flowing into an output capacitor suchthat a charge time of the capacitor is minimized when the current ismaximized.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a battery powered sensor node.

FIG. 2 is a block diagram of an energy harvesting sensor node.

FIG. 3 is a block diagram of an exemplary energy harvesting image sensornode.

FIG. 4 is a block diagram of an energy harvesting system powered byimage sensor pixels.

FIG. 5 is a block diagram of an output voltage regulator using aswitched capacitor (SC) boost converter with ping-pong hystereticcontrol.

FIG. 6 is a graphical illustration of control signals for the ping-ponghysteretic control of the SC boost converter shown in FIG. 5.

FIG. 7 is a block diagram of a non-overlap clock phase generator.

FIG. 8 is a graphical illustration of control and intermediate signalsof the non-overlap clock phase generator of FIG. 7.

FIG. 9 is a block diagram of a clock generator using interleavedoscillator with power gated buffers.

FIG. 10 is a block diagram of buffered high speed ring oscillator.

FIG. 11 is a block diagram of buffered high speed current starved ringoscillator.

FIG. 12A is a block diagram of a first exemplary non-overlap two phaseclock generator with a frequency independent delay.

FIG. 12B is a block diagram of a second exemplary non-overlap two phaseclock generator with a frequency independent delay.

FIG. 13 is a block diagram of a two phase clock generator withinterleaved ring oscillators.

FIG. 14 is a block diagram of interleaved current-starved ringoscillators with power-gated buffers.

FIG. 15 is a block diagram of interleaved current-starved ringoscillators with power-gated buffers and ungated buffers.

FIG. 16 is a block diagram of state machine based multiplenon-overlapping clock phase generator.

FIG. 17 is a block diagram of an energy harvesting system.

FIG. 18 is a block diagram of a maximum output power tracking regulatorcontrolling switching frequency or duty cycle.

FIG. 19 is a block diagram of an alternative embodiment of a maximuminput power tracking regulator controlling switching frequency or dutycycle.

FIG. 20 is a schematic diagram of a system model of a switchingfrequency regulator.

FIG. 21 is a block diagram of a regulator with a switched capacitorconverter, maximum output power tracking regulator, and ping-ponghysteretic controller.

FIG. 22 is a graphical representation of output voltage of a regulatorwith a switched capacitor converter and ping-pong hysteretic controlwith respect to time.

FIG. 23A is a schematic diagram of a hysteretic controller.

FIG. 23B is a graphical representation of output voltage and controlsignals with respect to time.

FIG. 24A is a schematic diagram of a ping-pong hysteretic controller.

FIG. 24B is a graphical representation of output voltage and controlsignals with respect to time.

FIG. 25 is flow diagram of a hill-climb algorithm to minimize outputcapacitor charge time.

FIG. 26 is a graphical representation of charge time and switchingfrequency with respect to time.

FIG. 27 is a block diagram of an asynchronous maximum output powertracking circuit.

FIG. 28 is a graphical representation of measured peak efficiency andtracking efficiency of a system using maximum power point tracking withrespect to time.

FIG. 29 is a perspective view of an image sensor with peripheral diodesconfigured to capture light adjacent to an image field of view.

FIG. 30 is block diagram of auxiliary energy harvesting diodesconfigured to power up a system via a DC-DC converter with a lowfrequency (LF) oscillator.

FIG. 31 is a block diagram of an image sensor in an energy harvestingmode using a DC-DC converter with a high frequency (HF) oscillator.

FIG. 32 is a diagram of a diode stack arranged adjacent to an imagesensor.

FIG. 33 is cross-sectional diagram of an embodiment of two semiconductorstructures configured to capture light.

FIG. 34 is cross-sectional diagram of an alternative embodiment of twosemiconductor structures configured to capture light.

FIG. 35 is a schematic diagram of image sensor node having peripheraldiodes to capture adjacent light used for cold start sequence ofself-powered energy harvester.

FIG. 36 is a schematic diagram of a comparator with a in-built referencethreshold for power-on-reset (POR).

FIG. 37 is a schematic diagram of a hysteretic comparator with areference voltage generated by an isolated photo diode.

FIG. 38 is a graphical representation of measured voltage and logiclevels of a cold start image sensor using peripheral diodes with respectto time.

FIG. 39 is a graphical representation of typical image sensorapplications based on an image capture rate with respect to time.

DETAILED DESCRIPTION

As required, detailed embodiments of the present invention are disclosedherein; however, it is to be understood that the disclosed embodimentsare merely exemplary of the invention that may be embodied in variousand alternative forms. The figures are not necessarily to scale; somefeatures may be exaggerated or minimized to show details of particularcomponents. Therefore, specific structural and functional detailsdisclosed herein are not to be interpreted as limiting, but merely as arepresentative basis for teaching one skilled in the art to variouslyemploy the present invention.

The term “substantially” may be used herein to describe disclosed orclaimed embodiments. The term “substantially” may modify a value orrelative characteristic disclosed or claimed in the present disclosure.In such instances, “substantially” may signify that the value orrelative characteristic it modifies is within ±0%, 0.1%, 0.5%, 1%, 2%,3%, 4%, 5% or 10% of the value or relative characteristic.

The use of distributed sensor nodes employing energy harvestingtechniques may improve the battery life and form factor. Using thesensor itself as an energy harvesting element is an attractive option tofurther reduce the cost and footprint of such nodes. A CMOS image sensoris one example where the pixel array can be reconfigured to harvestincident light. Several applications, including warehouse inventorytracking and structural fault detection, capture images at a slow ratesuch as once every 2-5 minutes and are thus compatible with ultra-lowenergy harvesting power levels. However, a key challenge to thiscircuitry configuration is to ensure a robust cold start and efficientoperation over a wide range of light intensities.

Energy harvesting from an imager pixel array utilizes the pixel array togenerate energy when not being used to capture an image. However, thepixel array needs modifications to multiplex between an energyharvesting mode and an image capture mode. The modifications result in alow fill factor and large pixel pitch (i.e., reduced image resolution).Moreover, the system modifications require a battery or an off-chipinductor for start-up (i.e., cold start). Also, for energy-efficientoperation under varying light intensity, prior-art designs employmaximum input power point tracking (MPPT). However, this does notnecessarily lead to maximum power delivery to the load.

Other related work in this area has explored more generic energyharvesters with power management capabilities. A dedicated charge-pumpis used to enable cold start, but this comes with significant areaoverhead. The design of employs off-chip solar cells, which add extracost and footprint to the sensor node.

FIG. 1 is a block diagram of a battery powered sensor node 100. Thesensor node 100 includes at least one sensor 102 such as an imagingsensor, microphone, humidity sensor, pressure sensor, infrared sensor,magnetic sensor, temperature sensor, or a combination thereof. An outputof the sensor 102 may then be converted from an analog signal to adigital signal via an Analog to Digital Converter (ADC) 104 and thedigital signal may then be processed by a Central Processing Unit (CPU)106. The sensor node also includes a communication node 108. Thecommunication node 108 may be a wireless node configured to communicatevia a wireless protocol such as cellular, 802.11 (Wi-Fi), 802.15,Optical (e.g., via an optical fiber), Infrared (e.g., IrDA), or soundwaves (e.g., ultrasonic). Here the sensor node 100 is powered by abattery 110 or alternatively connected to a power grid, for example viaan AC/DC converter.

FIG. 2 is a block diagram of an energy harvesting sensor node 200. Thesensor node 200 includes at least one sensor 202 such as an imagingsensor, microphone, humidity sensor, pressure sensor, infrared sensor,magnetic sensor, temperature sensor, or a combination thereof. An outputof the sensor 202 may then be converted from an analog signal to adigital signal via an Analog to Digital Converter (ADC) 204 and thedigital signal may then be processed by a Central Processing Unit (CPU)106. The sensor node also includes a communication node 208. Thecommunication node 208 may be a wireless node configured to communicatevia a wireless protocol such as cellular, 802.11 (Wi-Fi), 802.15,Optical (e.g., via an optical fiber), Infrared (e.g., IrDA), or soundwaves (e.g., ultrasonic). Here the sensor node 200 is powered by thesensor 202 that includes an energy harvesting system or alternativelycan be selectively configured as an energy harvesting system.

FIG. 3 is a block diagram of an exemplary energy harvesting sensor node300. The sensor node 300 includes an integrated circuit (IC) 302 thatincludes a pixel array 304 and a DC/DC converter 306. The IC 302 isshown configured to operate the pixel array 304 as either a voltagesource to power the DC/DC converter generating at least one voltagelevel, or in an imaging mode in which the digital signal of the capturedimage is sent to an image readout circuit 308 that may be used to createa remote image 310.

For example, consider a fully integrated energy harvester IC usingconventional 4-T image sensor pixels with no change in fill factor orpixel pitch arranged as a QVGA (320×240) pixel array. This image sensorbeing configured to operate in an imaging mode (I_(M)) or an energyharvesting mode (EH) refer back to FIG. 3. This design includes a DC/DCconverter 306 such as a switched-capacitor (S_(C)) boost converter thatsteps up the pixel array voltage (V_(IN)) to produce two output voltages(A_(VDD) and D_(VDD)) for a standard imager readout. The readoutcircuitry was not included monolithically in this prototype, but due tothe unaltered pixel structure, this integration should be relativelystraightforward.

Next we will describe the IC chip architecture and its key buildingblocks followed by circuit details and concluding with measured resultsfrom a prototype fabricated in a sub 500 nm CMOS Image Sensor (CIS)process, however the concepts disclosed in this application are notlimited to these technologies.

FIG. 4 is a block diagram of an energy harvesting image sensor system400. The sensor system 400 includes an image area 402 with animage/pixel array 404 and auxiliary photodiodes. The array 404 iscoupled with a DC/DC converter 406 and voltage regulator 408 in whichthe regulated output is feedback to the DC/DC converter 406 via a MPPTcontroller 401 and oscillator 412. The QVGA pixel array is forwardbiased when configured in Energy Harvesting (EH) mode. When illuminated,the pixel array 404 harvests energy and generates V_(IN) between0.25-0.4 V, depending on the light intensity. This voltage (V_(IN)) isboosted to digital (D_(VDD)) and analog (A_(VDD)) supplies, which areregulated to 0.6 V and 1.8 V using hysteretic control circuits. The SCconverter operates with non-overlapping clocks from an oscillator. Theoscillator frequency is set by an MPPT algorithm to maximize the powerdelivered to the load under varying light and load conditions. Theauxiliary photodiodes, adjacent to pixel array, capture fringe light atthe periphery of the imager pixel array to enable cold start.

The QVGA pixel array consists of conventional 4-T pixels. Each pixel isembedded in a shared PWell and deep NWell as shown in FIGS. 33 and 34.The PWell-to-Ndiffusion (PD1) junction is reverse biased while thePWell-to-deep-NWell (PD2) junction is shorted during Imagining (IM)mode. The PD1's junction depth is unaltered and therefore has anegligible effect on image quality.

During EH mode, the NMOS transfer gate and reset transistor of the 4-Tpixel biases the Ndiffusion to GND. The forward biased PD1 and PD2junctions harvest energy from light at different wavelengths, generatinga positive VIN at the PWell node. The full array may be configured toshare one set of reconfiguration switches outside the array to multiplexit between IM and EH mode. This approach is pixel-pitch agnostic anddoes not sacrifice fill factor. As a result, this work achieves a highreported pixel fill factor (60.4%) and small pitch (e.g., 5 μm) for EHpixels.

Each unit photodiode in the auxiliary array may have a structure similarto that in the pixel array and is placed in its own deep NWell. Thephoto diodes may be configured in series, parallel, or a combinationthereof. For example, the photodiodes may be arranged with 2, 3, 4, 5,6, 7, 8, 9, 10, 11, 12, 13, 14, or 15 in series creating a photo diodestring of a desired voltage which then may be in parallel with otherphoto diode strings to achieve the desired current. The test chip wasbuilt with nine photodiodes stacked in series (refer FIG. 32). Thisstack generates an unregulated auxiliary supply rail (V_(AUX)) that canreach about 1.8 V. The diodes are progressively sized (4× to 35×) tocompensate for reverse photocurrent loss (I_(s,leak)) at thedeep-NWell-PSubstrate junctions. This sizing greatly enhances the outputdrive capability of the V_(AUX) rail.

FIG. 5 is a block diagram of an output voltage regulator 500 usingswitched capacitor (SC) boost converters with ping-pong hystereticcontrol. The voltage regulator 500 uses a modified Dickson topology forfully integrated SC boost conversion. This topology enables stepcharging the capacitors to reduce the conduction loss across theswitches. The topology also ensures minimal voltage excursions acrossthe top- and bottom-plate parasitic capacitors to reduce losses. The SCBoost-1 and SC Boost-2 converters are daisy-chained to generate the(unregulated) V_(DIG) and V_(ANA) voltages in incremental steps. Theseoutputs are then regulated using hysteretic control to generate D_(VDD)and A_(VDD). SC Boost-1 operates on non-overlapping clocks (φ1 and φ2)at frequency f_(HF), which is provided by a high-frequency oscillator.As SC Boost-2 takes its input from SC Boost-1, it operates on a gatedclock with a lower frequency f_(DIV). MPPT control regulates f_(HF) andf_(DIV) to maximize the output power delivered by each stage. FIG. 6 isa graphical illustration of control signals for the SC boost convertersshown in FIG. 5.

The test device had a boost converter with 10 stages, each containing200 pF unit fly capacitors. The converter can be programmed to tap twooutputs from any two distinct stages, which provides the flexibility tovary the conversion ratios independently. The V_(DIG) output from SCBoost-1 is nominally tapped after stage 2 and the V_(ANA) output from SCBoost-2 is tapped after stage 6, refer to FIG. 5. The clock of unusedstages are turned off to conserve energy.

The voltage regulation occurs through ping-pong hysteretic control. Thetwo output capacitors C1 and C2 buffer the boost converter outputs inopposite phases. When one capacitor delivers energy to the load, theother buffers the boosted output. The charge enable signal QEN1 chargesC1, while the load-enable signal LEN2 connects C2 to the load. Likewise,QEN2 and LEN1 begin in-phase once both QEN1 and LEN2 have been disabled.Comparing the voltages across C1 and C2 to V_(refL) and V_(refH) allowsfor independent control of charge and discharge time.

Depending on the loads at A_(VDD) and D_(VDD), the switching frequencyf_(HF) typically ranges from 0.5 to 10 kHz, while the ratiof_(DIV)/f_(HF) varies from 0.75 to 1. The clock generators and driversfor SC boost converter switches are often a major source of energy lossin energy harvesters. This is addressed via an improved Interleaved RingOscillator (IRO) design described below.

The chip's oscillators operate between 82 Hz (LF for cold start; seebelow) and 10 kHz (HF). Here, an ultra-low power Interleaved RingOscillator (IRO) that generates two non-overlapping clock phases isdisclosed, refer to FIGS. 14 and 15. This IRO draws a current that isproportional to its frequency, owing to a special circuit configurationthat minimizes shoot-through currents in its proceeding buffers (whichwould otherwise lead to a disproportional overhead). The IRO employs twointerleaved ring oscillator chains working at the same frequency, butout of phase. Each chain is made up of odd numbered (e.g., five in thisembodiment) current-starved stages to control the oscillation frequency.The current sources in each stage are gated to control the phase ofcharge and discharge. Stage outputs 1-5 and 1′-5′ control the gatingcontrols in the other chain. This leads to i and i′ (1≤i≤5) being out ofphase. To minimize shoot through currents in the buffers following thering oscillators, specific i and i′ nodes are used for power gating. Sixgated buffer stages are used to restore small transition times andconsume mainly dynamic switching current. Conventional clock driversfollow the gated buffers and drive the boost converter switches.

The measured IRO current consumption of the test device is 2.1 nA/kHzwith negligible shoot-through current at the lowest operating frequencyof 82 Hz. The current sources that bias the ring oscillator set theoscillator frequency. The non-overlap delay is proportional to the riseand fall times of stage nodes, which relaxes the clock driverrequirements.

Here, circuit techniques for generating ultra-low power clock phases forsignal processing and power conversion are disclosed. The disclosuredescribes generating non-overlapping clocks over a very wide frequencyrange at very low power consumption.

Most signal processing or power conversion ICs use switched capacitor orinductor circuits, however in energy-constrained applications likeenergy harvesting devices, wireless senor nodes or implantable devices,one of the significant sources of energy consumption is generating thesenon-overlapping clocks. In these applications and others, the clockgenerator needs to operate over a wide range of frequencies depending onthe mode of operation or available power.

Conventional on-chip non-overlap clock generator is made of the blocksin the dashed rectangle followed by clock drivers indicated in FIGS. 7and 8.

FIG. 7 is a block diagram of a non-overlap clock phase generator 700.The clock generator 700 includes an oscillator 702, buffers 706, a twophase non-overlapping generator 708 and clock drivers 710. FIG. 8 is agraphical illustration of control and intermediate signals of thenon-overlap clock phase generator of FIG. 7.

FIG. 9 is a block diagram of a clock generator 900 with a bufferedtwo-phase interleaving oscillator 902. The buffered two-phaseinterleaving oscillator 902 includes an interleaved oscillator 904 withpower gated buffers 906. The buffered two-phase interleaving oscillator902 provides input to clock drivers 908.

FIG. 10 is a block diagram of buffered high speed ring oscillator 1000.The buffered high speed ring oscillator 1000 includes a ring oscillator1002, buffers 1004, and a non-overlapping generator 1006. The ringoscillator 1002 is shown as 5 stage ring oscillator, however the ringoscillator may be any odd number of stages, such as a 5, 7, 9, 11, 21,31, 41, 51, etc. Also, the buffers 1004 is shown as a single buffer,however this may be implemented as multiple buffers arranged in seriesor parallel or a combination thereof.

A current starved ring oscillator or relaxation oscillator is oftenemployed to generate variable frequency clocks at lower power. When theclock frequency is reduced beyond a certain limit, the rise time (T_(r))and fall time (T_(f)) of the oscillator output (ϕ_(OSC)) increases. Thisresults in higher shoot-through current in the buffers following theoscillator and any reduction in power obtained by reducing theoscillator frequency can be lost.

FIG. 11 is a block diagram of buffered current starved high speed ringoscillator 1100. The buffered high speed ring oscillator 1100 includes aring oscillator 1102, buffers 1104, and a non-overlapping generator1106. The ring oscillator 1102 is shown as 5 stage ring oscillator,however the ring oscillator may be any odd number of stages, such as a5, 7, 9, 11, 21, 31, 41, 51, etc. Also, the buffers 1104 is shown as asingle buffer, however this may be implemented as multiple buffersarranged in series or parallel or a combination thereof.

To reduce the number of clock drivers and repeaters in the chip, thenon-overlap period (T_(nov)) must be large when the input supply poweris limited. This non-overlap period depends on the delay introduced inthe non-overlap generator. Conventionally, increasing the length of thedelay also increases power consumption of the non-overlap generator.

To overcome these significant issues at lower supply powers, we proposea new interleaved oscillator to generate non-overlapping clocks (ϕA andϕB) without the need of extra non-overlap generator circuits. Thenon-overlap period increases as frequency reduces, thereby enablingreduced clock drivers in energy-constrained applications. To reduce therise and fall times of the clocks, we propose new power gated buffers toavoid the shoot through current.

A conventional ring oscillator has odd number of inverter stages. Theoscillator output is buffered before being driven by clock drivers. Toreduce the power through the oscillator, the inverters in the oscillatorare made weaker (smaller W/L size). This increases the rise and falltimes of the ring oscillator output causing large shoot-through currentin the buffer.

A current starved ring oscillator is used to tune the frequency using aprogrammable current source in series with the inverters. As shown inFIG. 11, the output of the current starved oscillator also has largerise and fall times to achieve low frequency clocks. However, at lowerfrequencies, the shoot-through current in the buffers following currentstarved ring oscillator dominate the total power consumption.

FIG. 12A is a block diagram of a first exemplary CMOS logic circuitnon-overlap two phase clock generator 1200 with a frequency independentdelay. FIG. 12B is a block diagram of a second exemplary CMOS logiccircuit non-overlap two phase clock generator 1250 with a frequencyindependent delay. Here the delay stages I_(A1) to I_(AN), I_(B1) toI_(BN), and I₁ to I_(N) may be any number of inverters or similar logicstructure.

For low power, low frequency applications, it is desirable to havelarger non-overlap phases. This enables using reduced number of bufferstages while tolerating larger clock skew. However, larger delays comeat the cost of power consumption in these delay stages.

For high frequency clock generated from the current starved ringoscillator, the rise and fall times are small and hence, smallernon-overlap phase is sufficient to ensure two-phase operation. Smallernon-overlap phase ensures larger on-time in high frequency applications.This improves transient settling behavior in switched capacitor orinductor circuits.

Hence, it would be desirable to have delay between the non-overlap clockphases proportional to its frequency.

FIG. 13 is a block diagram of a two phase clock generator 1300 withinterleaved ring oscillators 1302. The interleaved ring oscillator 1302includes a first current starved ring oscillator 1304 and a secondcurrent starved ring oscillator 1306 which are coupled such that theoutput of the first current starved ring oscillator 1304 is an input tothe second current starved ring oscillator 1306, and the second currentstarved ring oscillator 1306 is an input to the first current starvedring oscillator 1304. The first current starved ring oscillator 1304 andthe second current starved ring oscillator 1306 send a signal to powergated buffers 1308 and 1310. The output of the first current starvedring oscillator 1304 is an input to the first power gated buffer 1308and gating control for the first power gated buffer 1308 is the outputof the second current starved ring oscillator 1306. Similarly, theoutput of the second current starved ring oscillator 1306 is an input tothe second power gated buffer 1310 and gating control for the secondpower gated buffer 1310 is the output of the first current starved ringoscillator 1304. The time-period or clock frequency may be based on theprogrammable current source in the current starved inverters.

Interleaving ring oscillators help generate clock phases at half theoriginal phase difference. These intermediate clock phases are used asgating control signals to prevent the shoot-through current in thepower-gated buffers as shown in FIG. 13.

Once, the edges of clocks are sharp enough after a few stages ofpower-gated buffers, regular buffers or clock drivers help drive thenon-overlapping clocks.

This approach also inherently introduces delays between the clock phasesthat are also proportional to frequency. Thus, low-power low-frequencyoperations can be implemented using fewer clock repeater stages.

At higher available power, clock frequency can be increased depending onthe mode of operation with smaller non-overlap period. This feature helpimprove settling of switched capacitor voltages.

FIG. 14 is a block diagram of interleaved current-starved ringoscillators with power-gated buffers system 1400. The interleaved ringoscillator 1402 includes a first current starved ring oscillator 1404and a second current starved ring oscillator 1406 which are coupled suchthat the output of the 1^(st) stage first current starved ringoscillator 1404 gates the 1^(st) stage of second current starved ringoscillator 1406, and the 1^(st) stage of second current starved ringoscillator 1406 gates the 1^(st) stage of first current starved ringoscillator 1404. The third current starved ring oscillator 1404 and thesecond current starved ring oscillator 1406 send a signal to power gatedbuffers 1408 and 1410. The output of the first current starved ringoscillator 1404 is an input to the first power gated buffer 1408 andgating control for the first power gated buffer 1408 is the output ofthe second current starved ring oscillator 1406. Similarly, the outputof the second current starved ring oscillator 1406 is an input to thesecond power gated buffer 1410 and gating control for the second powergated buffer 1410 is the output of the first current starved ringoscillator 1404.

The ring oscillator 1404, 1406 are shown as 5 stage ring oscillator,however the ring oscillators may be any odd number of stages, such as a5, 7, 9, 11, 21, 31, 41, 51, etc. Also, the buffers 1108, 1410 are shownas a double buffer, however this may be implemented as multiple buffersarranged in series or parallel or a combination thereof.

Also, each internal stage of the first ring oscillator 1404 isinterleaved and is used to gate the internal stages of the second ringoscillator 1406, and each internal stage of the second ring oscillator1406 is interleaved and is used to gate the internal stages of the firstring oscillator 1404. For example, the output A₁ to the first inverterstage of the first ring oscillator I_(A1) it used to gate the firstinverter stage of the second ring oscillator I_(B1), and the output B₁to the first inverter stage of the second ring oscillator I_(B1) it usedto gate the first inverter stage of the first ring oscillator I_(A1).This continues with each stage of each ring oscillator.

The power consumed by the interleaved ring oscillator and power-gatedbuffers was seen to scale linearly with the clock frequency which isindicative that there is not shoot-through current even at very lowclock frequencies. Also, the power of the non-overlap clock phasegenerator was seen to scales linearly with supply voltage.

FIG. 14 shows two interleaved current starved ring oscillators each with5 stages. The current sources in each stage help control the frequency.

Each of the current-starved inverters in the ring oscillator-A is gatedby the switching outputs (B1-B5) of corresponding inverters inoscillator-B respectively. Likewise, inverters in oscillator-B are gatedby switching outputs (A1-A5) from oscillator-A respectively.

These interleaving ring oscillators produce clock phases that are 18°out of phase of each other and create a output 180 degrees out of phase.The outputs from these oscillators have same frequency.

Taking the outputs from nodes A5 and B5, we get non-overlapping clocksin opposite phases. However, the rise and fall times are very largetransition times at low clock frequencies.

These large transition times would have led to large shoot throughcurrent in subsequent buffers. We resolved this issue using power-gatedbuffers. The power-gating signal for the buffers was selected to be oneof suitable oscillator phases A1-A5 and B1-B5 already generated. This isshown in FIG. 14 in which the first power gated buffer B_(A1) is gatedby the output of the second inverter stage of the first ring oscillatorI_(A2), and the first power gated buffer B_(B1) is gated by the outputof the second inverter stage of the second ring oscillator I_(B2). Inthis example, a second power gated buffer was used in which the secondpower gated buffer B_(A2) is gated by the output of the third inverterstage of the first ring oscillator I_(A3), and the second power gatedbuffer B_(B2) is gated by the output of the third inverter stage of thesecond ring oscillator I_(B3).

As an example, if A5 is rising, the output of the first inverterfollowing it is going to transition from high→low. Hence, to avoidshoot-through current in that inverter, it is gated by clock phase B2.This leads to outputs with lower transition times without anyshoot-through current penalty.

Generating sharper clock phases in this way leads to clocks withnon-overlap period that is proportional to the slow rise and fall timesof the interleaved ring oscillator. At lower supply voltage levels andlower clock frequency, we obtain larger non-overlap period. On the otherhand, at faster clock frequency, the non-overlap period is reduced.

FIG. 15 is a block diagram of interleaved current-starved ringoscillators with power-gated buffers and ungated buffers system 1500.

FIG. 15 is a block diagram of interleaved current-starved ringoscillators with power-gated buffers system 1500. The interleaved ringoscillator 1502 includes a first current starved ring oscillator 1504and a second current starved ring oscillator 1506 which are coupled suchthat the output of the 1^(st) stage first current starved ringoscillator 1504 gates the 1^(st) stage second current starved ringoscillator 1506, and the 1^(st) stage second current starved ringoscillator 1506 gates the 1^(st) stage first current starved ringoscillator 1504. The third current starved ring oscillator 1504 and thesecond current starved ring oscillator 1506 send a signal to power gatedbuffers 1508 and 1510. The output of the first current starved ringoscillator 1504 is an input to the first power gated buffer 1508 andgating control for the first power gated buffer 1508 is the output ofthe second current starved ring oscillator 1506. Similarly, the outputof the second current starved ring oscillator 1506 is an input to thesecond power gated buffer 1510 and gating control for the second powergated buffer 1510 is the output of the first current starved ringoscillator 1504.

Regular clock driver buffers increase the driving strength of thesenon-overlapping clocks. A simple state machine with multiplexers to passone of the two clocks can generate any number of non-overlapping clockphases as shown in FIG. 16.

FIG. 16 is a block diagram of state machine based multiplenon-overlapping clock phase generator 1600. The clock generator includesa state machine 1602 and multiplexers. In this exemplary block diagram,there are three multiplexers (a first multiplexer 1604, a secondmultiplexer 1606, and a third multiplexer 1608) used to create a threephase clock. However more or less multiplexer can be used to generate adesired number of phases. For example, if only two phases are neededthen only a first multiplexer 1604, a second multiplexer 1606 would beneeded and the third multiplexer 1608 could be removed. If more phasesare needed then additional multiplexers could be added.

An energy harvester has a transducer that converts physical energy likevibration, light or temperature difference to electrical energy. Thiselectrical energy is often generated at a voltage that is different fromthe voltage required to charge the battery or a capacitor. A switchingvoltage converter is generally used to step up/down the voltage.

Tracking maximum output power under varying inputs is challenging inenergy constrained applications. This disclosure outlines an analogmaximum output power tracking scheme that regulates the switchingfrequency of the voltage regulator using hill climbing (also known asperturb and observe) algorithm at a very low energy. The proposed methodadjusts the switching frequency to vary the input and output impedancesof the voltage regulator. This impedance matching helps deliver maximumpower output from the harvester.

Energy harvesting systems generally regulate output voltage withhysteretic control due to its simplicity and low power implementation.This disclosure describes an improved modified hysteretic control withhigher efficiency that also provide the control signals for trackingoutput power.

A typical energy harvesting system has a transducer that convertsphysical energy like light, pressure or heat to electrical energy. Theelectrical energy from the transducer is produced at a voltage V_(IN).This needs to be stepped up/down before it can be delivered to the loadR_(LOAD) or stored in a capacitor C_(LOAD).

FIG. 17 is a block diagram of an energy harvesting system 1700. Theenergy harvesting system 1700 includes a clock generator 1702, atransducer 1704, a switching voltage converter 1706, a voltage regulator1708, and a load having a resistance load R_(LOAD) and/or a capacitanceC_(LOAD). The transducer 1704 includes an imaging sensor, microphone,humidity sensor, pressure sensor, infrared sensor, magnetic sensor,temperature sensor, or a combination thereof.

A switching voltage converter delivers output power regulated at voltageV_(OUT). The converter needs clocks to turn its switches on and offperiodically. Changing the switching frequency or duty cycle regulatesthe power delivered to the load from the source. A feedback loop sensesthe output power and modulates the switching frequency or duty cycle asshown in FIG. 18.

FIG. 18 is a block diagram of an energy harvesting system 1800 with aswitching frequency regulator. The energy harvesting system 1800includes a clock generator 1802, a transducer 1804, a switching voltageconverter 1806, a voltage regulator 1808, a load having a loadresistance R_(LOAD) and/or a load capacitance C_(LOAD), and a Maximumoutput power tracking module 1810. The transducer 1804 includes animaging sensor, microphone, humidity sensor, pressure sensor, infraredsensor, magnetic sensor, temperature sensor, or a combination thereof.

Sensing output power requires sensing output voltage and current andgetting its product to adjust the switching frequency. This is extremelychallenging in energy constrained applications like energy harvestingfrom ambient sources.

A way to maximize power in an energy harvesting system is shown in FIG.19. FIG. 19 is a block diagram of an energy harvesting system 1900 witha switching frequency regulator. The energy harvesting system 1900includes a clock generator 1902, a transducer 1904, a switching voltageconverter 1906, a voltage regulator 1908, a load having a loadresistance R_(LOAD) and/or a load capacitance C_(LOAD), and a Maximumoutput power tracking module 1910. The transducer 1904 includes animaging sensor, microphone, humidity sensor, pressure sensor, infraredsensor, magnetic sensor, temperature sensor, or a combination thereof.

Each source has a definite characteristic in its region of operationwhere its power can be maximized. For example, a thermoelectricgenerator (TEG) delivers maximum power when the voltage across it ishalf the voltage under no-load condition. Likewise, a photoelectric cellor solar cell delivers maximum power when the voltage across it isaround 0.8 times the voltage under no-load condition.

A simple and relatively low-power implementation of measuring the sourcevoltage under no-load condition leads to the feedback control systemshown in FIG. 19. Regulating the voltage (V_(IN)) delivered by thesource by controlling the switching frequency or duty cycle enablesmaximum power extraction from the source.

The functioning of this system can be better described from the overallsystem model shown in FIG. 20. FIG. 20 is a schematic diagram of asystem model 2000 of a switching frequency regulator. The transducersource 2002 can be modelled as a Thevenin voltage source V_(S) with anequivalent impedance R_(S). The switching voltage converter 2004 ismodelled as a voltage converter with a ratio 1:N and an equivalentoutput impedance that is inversely proportional to switching frequencyf_(CLK). And the load 2006 is modeled as a load resistance R_(LOAD) anda load capacitance C_(LOAD).

Controlling the frequency or duty cycle of the clock, the impedancelooking into the converter (R_(IN)) is modulated such that V_(IN)becomes some fraction of V_(S) depending on the energy source. Undersuch a condition, R_(IN) becomes equal to R_(S) and maximum power isextracted from the energy source. Thus, optimizing the clock to haveimpedance matching in matching network-1 lead to sub-optimal operationof matching network-2. This results in lower overall power output fromthe voltage converter.

This disclosure relates to regulating the clock to maximize the overallpower output from the voltage converter. The primary advantage of thisscheme is that it enables maximum power delivered from an energyharvesting system and is not limited to extraction of maximum power fromthe source. A few novel points include:

Some key aspects of this maximum power tracking system include:

This system can measure and maximize the output power in energyconstrained applications like energy harvesting, yet the implementationis not limited to maximizing power from the energy source.

A method to sense output power by measuring current from an outputvoltage regulator while consuming very low energy is achieved.

The current measurement from voltage regulator does not need a seriesresistor or a replica generation. The current output from the voltageregulator is measured by measuring the charging time of an intermediatecapacitor in the voltage regulator.

The charging time of the intermediate capacitor is measured byconverting a small bias current to a corresponding voltage and using avoltage regulator to compare the charging times at different clocks.

A new multiplexed ping-pong hysteretic output voltage regulator schemeis disclosed that improves the efficiency but maintains the simplicityof conventional hysteretic control.

FIG. 21 is a block diagram of a system 2100 with a regulator having aswitched capacitor converter and ping-pong hysteretic control. Theenergy harvesting system 2100 includes a clock generator 2102, atransducer 2104, a switching voltage converter 2106 such as a switchedcapacitor converter, a ping-pong hysteretic control voltage regulator2108, a load having a load resistance R_(LOAD) and/or a load capacitanceC_(LOAD), and a Maximum output power tracking module 2110. Thetransducer 2104 includes an imaging sensor, microphone, humidity sensor,pressure sensor, infrared sensor, magnetic sensor, temperature sensor,or a combination thereof.

In one embodiment, the system 2100 is a fully integrated switchedcapacitor converter with hysteretic control. The system may regulate theoutput power by changing the switching frequency of the switchedcapacitor converter 2106. The output voltage is regulated using a newping-pong hysteretic controller 2108 which can improve the efficiency inthe voltage regulator. The techniques described here are not limited tothe type of voltage regulator. The feedback control variable is also notlimited to the switching frequency and can include duty cycle or anyother variable with which the voltage regulator's impedance can becontrolled. The output power is a product of the output voltage andcurrent which can be represented as equation 1.Pout=Vout*Iout  (1)

The ability to sense an output voltage and an output current todetermine the product is challenging for low energy systems. The outputvoltage is regulated within a hysteretic band as shown in FIG. 22. FIG.22 is a graphical representation 2200 of output voltage 2202 of aregulator with a switched capacitor converter and ping-pong hystereticcontrol with respect to time 2204. In this graphical representation2200, the output voltage waveform 2206 ping-pongs between high outputvoltage at time 2208 and a low output voltage at time 2210 with a periodbetween times 2208 and 2212.

The ripple voltage on V_(OUT) (=V_(OUT,HIGH)−V_(OUT,LOW)) is designed tobe small, with the output voltage remaining mostly constant and hence,the output power is proportional to output current. To maximize outputpower, the output current from the converter needs to be maximized.

To sense the output current, a current is measured by measuring thevoltage across a series resistor or using a replica current mirror.However, these require additional circuits and consume power to sensethe current.

Here, the use of a time to charge an output capacitor in the hystereticcontroller is used as an estimate of the output current as illustratedin FIG. 23A. FIG. 23A is a schematic diagram 2300 of a hystereticcontroller using a Schmidt trigger 2302. FIG. 23B is a graphicalrepresentation 2350 of output voltage 2352 and control signals 2354,2356 with respect to time 2358. Higher the charging current, lower isthe charging time.

The output current from the converter charges the output capacitor C_(O)when QEN (charge-enable) is high. C_(O) discharges to deliver power tothe load when QEN is low. When delivering power to the load, the energyfrom the disconnected converter is not utilized leading to inefficientoperation.

To rectify this issue, a ping-pong hysteretic control technique isdisclosed in FIG. 34A. FIG. 24A is a schematic diagram of a ping-ponghysteretic controller 2400 which includes combination logic 2402 togenerate control signals. FIG. 24B is a graphical representation 2450 ofoutput voltage 2452 and control signals 2454, 2456, 2458, and 2460 withrespect to time 2462. The use of two capacitors stores the output fromthe converter before it is delivered to the load. As a first capacitorC_(O1) is charging from the converter, the second capacitor C_(O2)discharges to deliver power to the load. Comparing each node to eitherV_(OUT,HIGH) or V_(OUT,LOW) gives independent control of charging time.

This ping-pong of two capacitors to deliver output power without wastingconverter's energy during load delivery makes it more efficient thanconventional hysteretic control. In other embodiments, 3 or morecapacitors can be used in which the charge is provided to each capacitorin a round robin fashion.

To measure the output current from the converter, the charging time inone or both capacitors can be observed. A hill climbing or perturb andobserve algorithm is used to adjust the switching frequency of the clockin such a way that the charging time of the output capacitors isminimized. This algorithm is explained with the flowchart in FIG. 25.

FIG. 25 is flow diagram of a hill-climb algorithm 2500 to minimizeoutput capacitor charge time. Here a controller begins operation at step2502. At step 2504, the controller initializes a minimum charge time toan initial value such as a maximum value based on the capacitors valuesand output load conditions. At step 2506, the controller decreases theswitching frequency (f_(CLK)), the decrease amount may be a fixedamount, or a variable amount. At step 2508, the controller monitors thecharge time.

At step 2510, the controller compares the charge time with a minimumcharge time. If the charge time is less than the minimum charge time thecontroller sets the minimum charge time to the current charge time instep 2512 and branches back to step 2506. If the charge time is greaterthan or equal to the minimum charge time the controller branches to step2514. At step 2514, the controller increases the switching frequency(f_(CLK)), and proceeds to step 2516 in which the controller monitorsthe charging time.

At step 2518, the controller compares the charge time with a minimumcharge time. If the charge time is less than the minimum charge time thecontroller sets the minimum charge time to the current charge time instep 2520 and branches back to step 2514. If the charge time is greaterthan or equal to the minimum charge time the controller branches to step2522. At step 2522, the controller decreases the switching frequency(f_(CLK)) to the previous value and exits the loop at step 2524.

FIG. 26 is a graphical representation 2600 of charge time 2602 andswitching frequency 2604 with respect to time 2606. The variation inswitching frequency over a period of time are shown in FIG. 26 for oneexample case. The algorithm reduces switching frequency from f1 to f2and the observed charging time increases. Thus, the frequency updatedirection is reversed and we keep increasing the switching frequency f2to f5 and observe that charging time keeps reducing. This shows that weare operating our complete system to deliver more output power from theconverter. If we increase the switching frequency further to f6, thecharging time increase and we stop the hill-climbing algorithm. Weselect the penultimate value of the switching frequency (f5 in our case)as the most optimal.

The charge time is sensed by converting it to a proportional voltage sothat it can be then compared using conventional voltage comparators. Asmall bias current I_(BIAS) is used to charge capacitors (C_(X) orC_(Y)) for charge time as shown in FIG. 27.

FIG. 27 is a block diagram of an asynchronous maximum output powertracking circuit 2700. The asynchronous maximum output power trackingcircuit 2700 includes a capacitor bank 2702 a Schmidt trigger comparator2704, hill climb logic 2706 that may be used to control the charging andresetting of capacitors in the capacitor bank 2702. The hill climb logic2706 feeds an up/down counter 2708 that then has a bias control 2710, toset a frequency of an at least one oscillator 2712.

Initially C_(X) is charged to supply V_(AUX) while C_(Y) stores thecharge time corresponding to current frequency f1. The frequency isreduced from f1 to f2 and the I_(BIAS) charges C_(X) for the increasedcharge time. Thus, the voltage comparator decides V_(X) to be greaterthan V_(Y). We keep the minimum value in C_(Y) and reset C_(X). Thehill-climbing logic asserts to increment the frequency and an up/downcounter sets the bias control for new switching frequency f3. I_(BIAS)recharges C_(X) for reduced charging time, in our example case, and thevoltage comparator decides that V_(X) is less than V_(Y). C_(Y) is nowreset and frequency update continues until charging time startsincreasing again.

The circuits proposed here can be designed to be asynchronous circuitsthat switch only when one charge period ends. This prevents powerdissipation at each clock.

Maximum Power Point Tracking (MPPT) control regulates the boostconverter clock frequency generated by the IRO. The boost converter'sswitching frequency modulates the input and output impedances of theconverter. Interfacing with finite source and load impedances, the MPPTcontrol maximizes load current delivery at the regulated outputvoltages. Minimizing the QEN1/QEN2 pulse width maximizes load currentand hence the output power delivered to the load.

In the circuit of FIG. 27, a small bias current (IBIAs) charges acapacitor (C_(X) or C_(Y)) and converts QEN1's charging time to voltage.Comparing V_(X) and V_(Y) helps compare the charging times at differentfrequencies. V_(X) is initialized to V_(AUX) and V_(Y) is initialized toGND during power up. The hill climbing logic block selects CA or CB toretain the smaller voltage and overwrite the larger voltage afterresetting it. Thus, the switching frequency progressively decreases orincreases until QEN1 pulse width starts increasing. This is establishedusing an increment or decrement signal to an up/down counter thatcontrols the biasing of the HF oscillator again referring to FIG. 27.

The hill climbing logic is implemented as a custom event-triggered logicthat evaluates the new states only on QEN1's falling edge. This slowevaluation enables ultra-low power operation and gives sufficient timefor updating the HF oscillator frequency. Thus, each QEN1 pulse widthcorresponds to a different switching frequency.

FIG. 28 is a graphical representation 2800 of measured peak efficiency2802 and tracking efficiency of a system using maximum power point 2804with respect to time 2806.

FIG. 28 shows the maximum boost converter efficiency and the trackingefficiency under various light intensities. The measurements were takenwith a 2-12 MΩ load on D_(VDD) and a 27-530 MΩ load on AV_(DD) (lowerload resistance at brighter illumination). The maximum boost converterefficiency is found by sweeping the switching frequency around theMPPT-predicted value. The MPPT tracking efficiency is the ratio of theefficiency predicted by the MPPT loop and the maximum efficiency foundby sweeping. The boost converter may work at a maximum efficiency of52.4% in indoor lighting condition of 430 lux. At a low light of 52 lux,the boost converter efficiency is 38.2% at a minimum input power of 79.1nW. The MPPT output power tracking efficiency exceeds 96%.

For a self-powered transducer such as an image sensor, the power supplyrails must be charged up to the required voltage before the image sensorcan start capturing images. This is accomplished using a DC-DC converterto generate these supply rails from incident light. However, operatingthis DC-DC converter also requires stable supply rails.

This disclosure addresses the cold-start of a DC-DC converter usingfringe incident light proximate to the image sensor. The disclosureillustrates how to convert very low levels of light to higher voltagesand establishing stable power supply rails after which a conventionalDC-DC converter can maintain the power supply rails and sensor can thenbe used to capture an image.

Light from the object passes through a lens or system of lenses andfalls on the image sensor. Some fraction of the light passing throughthe lens or system of lenses also falls on the peripheral area aroundthe image sensor. This fringe light energy is used to cold-start theDC-DC converter to generate the imager's power supply rails.

In this disclosure, the peripheral area is populated with photodiodesarranged in series, parallel, and a combination thereof configured toharvest an unregulated voltage, Vaux, which is high enough to allowoperation of circuitry but it has very limited current drive capacity.The auxiliary voltage (Vaux) enables/powers the operation of a lowfrequency (LF) oscillator which drives a DC-DC converter used togenerate the main supply voltage.

FIG. 29 is a perspective view of an integrated circuit (IC) 2902including an image sensor 2904 and a peripheral area 2906 having diodes.The image sensor is configured to capture light from an object gatheredvia a lens 2908 or lens system. The peripheral area 2906 is configuredto capture peripheral light 2910 passing through the lens 2908 incidentto the IC 2902 and adjacent to an area of the image sensor 2904.

FIG. 30 is block diagram of an auxiliary energy harvesting system 3000having auxiliary diodes 3008 proximate to an image sensor 3004 andconfigured to provide power via a DC-DC converter 3006 with a LFoscillator 3002.

FIG. 31 is a block diagram of an auxiliary energy harvesting system 3100having an image sensor 3104 in an energy harvesting mode using a DC-DCconverter 3106 clocked by a HF oscillator 3102 to feed a voltageregulator 3108 to generate a supply voltage V_(sup).

The primary input voltage (energy) to the DC-DC converter 3006 isgenerated by the image sensor 3004 configured as energy harvester. Usingthe LF oscillator 3002 the energy from Vaux is used to boost the voltageon Vsup to a sufficient level to allow a secondary high frequency (HF)oscillator 3102 to take over. Once in this regular operationconfiguration the cold-start has completed and the energy can be storedor used to drive a load (R_(load)) as shown in FIG. 31. Regularoperation can generate much more energy from the imager sensor 3004,3104 because it used a high frequency oscillator 3102 to drive the DC-DCconverter 3006, 3106.

During start up from complete power off, light first falls on the imagesensor, while all the power supply voltage rails are substantially at aground potential. Once these rails reach a threshold voltage level,regular conversion using a HF oscillator may begin as shown in FIG. 31.

FIG. 32 is a diagram of an imaging system 3200 including an integratedcircuit 3200 with a pixel array 3204 and a diode stack 3206 arrangedadjacent to the pixel array 3204. The diode stack has a first diode D₁in which the anode is a P-well (PW) and the cathode is a deep N-well(DNW), a second diode D₂ in which the anode is a P-well (PW) and thecathode is a deep N-well (DNW) and in series with the D₁, and forming astring of series diodes up to an N^(th) diode D_(n) in which the anodeis a P-well (PW) and the cathode is a deep N-well (DNW). The currentthrough D₁ is I₁ and the current through D₂ is I₂ which equals thecurrent through I₁ and a leakage current I_(R1) through the substratediode D_(R1). Also, the current through D_(n) is I_(n) which equals thecurrent through I₂ (I₁+I_(R1)) and a leakage current I_(R2) through thesubstrate diode D_(R2)

One embodiment of an arrangement of auxiliary photodiode stack aroundthe energy harvesting (EH) image sensor is shown in FIG. 32 that isbased on a test IC. The test IC was configured with nine photodiodesstacked to form a series connection. In other embodiments, the auxiliarydiode arrangement may not be around the complete circumference of thepixel array, but may be on a side of the pixel array, or multiple sidesof the pixel array. Also, the test chip combined both the pixel arrayand auxiliary diodes monolithically, however in other embodiments, theauxiliary diodes may be on a separate chip such that the two chips arecombined in a multi-chip module (MCM).

FIG. 33 is cross-sectional diagram 3300 of an embodiment of twomonolithic semiconductor structures, S₀ and S₁ configured to capturelight. A first structure S₀ has three junctions, J1 ₁ between anN-region and a P-well (PW), J2 ₁ between the PW and deep N-well (DNW),and J3 ₁ between a P substrate (PSUB) and the DNW. The three junctionform three diode D₀₁, D₀₂, D₀₃. This structure includes interconnectssuch that D₀₃ and D₀₂ are in parallel and D₀₁ is reverse biased.

FIG. 34 is cross-sectional diagram 3400 of an alternative embodiment oftwo semiconductor structures, S₀ and S₁ configured to capture light.Similar to FIG. 33, this structure has a first structure S₀ has threejunctions, J1 ₁ between an N-region and a P-well (PW), J2 ₁ between thePW and deep N-well (DNW), and J3 ₁ between a P substrate (PSUB) and theDNW. The three junction form three diode D₀₁, D₀₂, D₀₃. This structureincludes interconnects such that D₀₁ is forward biased and D₀₂ and D₀₁are reverse biased.

In the test chip, each photodiode in the stack was in a dedicatedseparate deep Nwell as shown in the cross-section of the structure inFIGS. 33 and 34. To account for the reverse saturation current leakageat each deep Nwell-P-substrate junction (I2, I3), the photodiodes lowerin the stack are made larger. The size of the photodiodes decreaseprogressively as we go higher in the stack as shown in the schematic ofFIG. 32. The lowest photodiode is larger as it has to carry larger(I1+I2+I3) current than the one on the above it which carries (I1+I2)current. This pyramid structure of entire stack of nine photodiodes isshown in the schematic of FIG. 32 and greatly improves the current drivecapability of generated Vaux supply range.

The sizing of the diodes is approximately in geometric progressionrounded to nearest integer. Each photodiode in the stack is made ofgroup of unit cells. This makes the layout of the stack in theperipheral area highly flexible while maintaining the geometricprogression ratio.

FIG. 35 is a schematic diagram of image sensor node 3500 havingperipheral diodes 3502 and an image sensor 3504. The peripheral diodes3502 are configured to harvest energy from light at the periphery to theimage sensor 3504. The image sensor 3504 is configured to harvest energyand to capture images. The system 3500 uses a low frequency (LF)oscillator 3506 to a DC-DC converter 3508 during cold start and thenswitches to a high frequency (HF) oscillator during normal operationonce sufficient power is achieved.

However, power sequencing must follow a well-defined order to preventany unwanted leakage through the chip and ensure proper functionality.This is sequence of steps for cold start are shown in FIG. 35.

For energy-autonomous operation, an IC was built such that the system3500 starts using the V_(AUX) supply rail when light first shines on thechip. The test device delivered approximately 25 nW at 1.8 V under dimindoor lighting condition (25 lux). The four-phase cold start sequenceshown in FIG. 35 is described as follows:

First the entire chip 3500 is power gated until the V_(AUX) ramp crossesa certain threshold and asserts the V_(AUXGD) (V_(AUX_GOOD)) signal.

Second, a comparator powered by V_(AUX) asserts V_(INGD) (V_(IN_GOOD))when V_(IN) exceeds a fraction of the open-circuit voltage (V_(OC)) ofan isolated pixel. In other words, when V_(IN) exceeds a thresholdlevel.

Third, an ultra-low power, low-frequency (LF) oscillator 3506, workingfrom the V_(AUX) rail, generates non-overlapping clocks for the SC boostconverter.

Fourth, the SC boost converter steps up V_(IN) to the A_(VDD) andD_(VDD) rails without delivering power to the load. Once the outputrails are within a regulation range (e.g., 1%, 2%, 3%, 4%, 5%, of adesired output voltage) the COLDST (cold start done) signal is assertedto begin normal operation with the HF oscillator and the same boostconverters. The always-on wake-up comparator A_(IS) in FIG. 35 may havea built in weight or offset to realize a built-in reference. WhenV_(AUX) exceeds the threshold (˜1.8 V) generated by the offset,V_(AUXGD) is asserted. The V_(AUXGD) dependent PMOS load deviceintroduces hysteresis.

A more detailed flow follow the below four steps:

Step 1: With incident light, Vaux from photodiode stack ramps to ahigher voltage. When it crosses a certain minimum threshold, Vaux_Goodsignal is asserted. This turns on the power gating switch and power fromVaux_Gated reaches all other parts of the chip. Until Vaux_Good isasserted, the rest of the chip does not get any power and hence, theonly power consuming block in the entire chip is ultra low-power Vauxcomparator.

Step 2: After Vaux_Good is asserted, input voltage from Vin from theimager pixels is compared to a certain reference. Once Vin reaches astable voltage above a certain minimum value, Vin_Good is asserted.

Step 3: Assertion of both Vaux_Good and Vin_Good implies that we canstart running the bias currents and LF oscillator from Vaux_Gated supplyat a very small current and very low frequencies.

Step 4: These low frequency clocks then drive the DC-DC converter togenerate the output Vsup voltage. Vsup reaching a minimum defined valueasserts Cold-Start Done signal. This assertion then enables regular highefficiency mode of conversion using HF oscillator.

Vin keeps getting converted to Vsup at a much higher efficiency andoutput load delivery is enabled. Output Vsup is regulated to be within acertain voltage band

An advantage of or this system and method is that it enables cold-startof a self-powered image sensor. This is achieved this without using anyextra battery, super capacitor, inductor, or charge pump in the circuitand therefore offers a cost and size advantage.

Each photodiodes have two vertically stacked light harvesting PNjunction: NDiff-Pwell and Pwell-deep Nwell. Each photodiode sits in itsown deep Nwell.

There is reverse photocurrent loss at each deep Nwell-Psubstratejunction shown in FIGS. 32, 33, and 34. In another embodiment, thephotodiodes in the stack are sized approximately in geometricprogression to account for the reverse current junction leakage betweentwo consecutive photodiodes. This enables higher current delivery fromVaux supply.

The ratio of the sizes of photodiodes is selected to be <2 to supply thedeep Nwell-Psub photocurrent loss while still keeping area of theauxiliary photodiode stack small enough to fit in the periphery of thepixel array.

Several small unit photodiodes are arranged in parallel to generate alarger photodiode. The smaller photodiodes can be arranged in theperipheral ring outside the pixel array shown in FIG. 32.

This cold-start scheme ensures each supply rail reaches its definedvoltage level without any leakage current or loss of functionality underall lighting conditions.

When the light is first incident on the chip, there is no referencevoltage to compare to Vaux, so a new skewed comparator with offset thatcan act as a reference is disclosed in FIG. 36.

And, a fraction of open circuit voltage (Voc) of an isolated photodiodeis used as a reference to compare Vin with is disclosed in FIG. 37.

The cold-start sequence shown in FIG. 35 begins with Vaux generationhaving the following four steps.

Step 1: Vaux is the supply and the input of a skewed comparator as shownin FIG. 36. FIG. 36 is a schematic diagram of comparator 3600 within-built reference threshold for power-on-reset (POR). When light firstfalls on photodiodes, Vaux is close to ground and I_(L)˜I_(R)˜0.Initially with small Vaux, I_(R)>I_(L) due to the skewed sizing.However, as Vaux ramps up beyond a certain voltage level, I_(L)>I_(R)due to the exponential dependence of drain current of transistor on gatevoltage when operating in subthreshold region. In this way, referencevoltage is inbuilt into the comparator.

Ultra-low power operation is achieved with tail current source biased indeep subthreshold. Hysteresis in the threshold is introduced by skewingthe load of the comparator depending on current Vaux_Good value. In thetest chip to provide an offset in the comparator for the steps, theV_(IN1) was set to be 3*V_(AUX)/8, V_(IN2) was set to be 5*V_(AUX)/8,and V_(BIAS) was set to be V_(AUX)/8 such that0<V_(BIAS)<I_(IN1)<V_(IN2)<V_(AUX).

Step 2: When Vaux_Good is asserted, the power-gated Vaux_Gated supplyenables the comparators to check if the V_(IN) rail from the imagerpixels is high enough to be loaded. The reference for the Vin_Goodgeneration is implemented using a fraction of open circuit voltage (Voc)of an isolated photodiode as illustrated in FIG. 37. Voc is the maximumvoltage attained by unloaded Vin.

FIG. 37 is a schematic diagram 3700 of hysteretic comparator withreference voltage generated by an isolated photo diode D₁. A firstcomparator 3702 and second comparator 3704 have a reference voltageinput from a voltage divider 3708 that is used to create referencevoltages from the diode D₁ voltage. The output of the first comparator3702 is used to enable the Flip-Flop, while the output from the secondcomparator 3704 is used to drive the active low RESET of the Flip-Flop.

Step 3: After Vaux_Good and Vin_Good assertion, low power current biasgeneration followed by non-overlapping clock generation using LFoscillator is implemented.

Step 4: These non-overlapping clocks drive the DC-DC converter togenerate stable Vsup power rail. Once this rail is established, regularoperation using HF oscillator delivers power to the load at regulatedvoltage level.

FIG. 38 is a graphical representation 3800 of measured voltage 3802 andlogic levels 3804 of a cold start image sensor using peripheral diodeswith respect to time.

FIG. 38 illustrates the measured cold start sequence waveforms.V_(AUXGD) is asserted once V_(AUX) ramps above 1.8 V and V_(INGD) isasserted when V_(IN) exceeds 0.7 V_(OC). Once DV_(DDGD) and AV_(DDGD)are asserted, the chip begins normal operation and provides regulatedoutput voltages.

The program code embodying the algorithms and/or methodologies describedherein is capable of being individually or collectively distributed as aprogram product in a variety of different forms. The program code may bedistributed using a computer readable storage medium having computerreadable program instructions thereon for causing a processor to carryout aspects of one or more embodiments. Computer readable storage media,which is inherently non-transitory, may include volatile andnon-volatile, and removable and non-removable tangible media implementedin any method or technology for storage of information, such ascomputer-readable instructions, data structures, program modules, orother data. Computer readable storage media may further include RAM,ROM, erasable programmable read-only memory (EPROM), electricallyerasable programmable read-only memory (EEPROM), flash memory or othersolid state memory technology, portable compact disc read-only memory(CD-ROM), or other optical storage, magnetic cassettes, magnetic tape,magnetic disk storage or other magnetic storage devices, or any othermedium that can be used to store the desired information and which canbe read by a computer. Computer readable program instructions may bedownloaded to a computer, another type of programmable data processingapparatus, or another device from a computer readable storage medium orto an external computer or external storage device via a network.

Computer readable program instructions stored in a computer readablemedium may be used to direct a computer, other types of programmabledata processing apparatus, or other devices to function in a particularmanner, such that the instructions stored in the computer readablemedium produce an article of manufacture including instructions thatimplement the functions, acts, and/or operations specified in theflowcharts or diagrams. In certain alternative embodiments, thefunctions, acts, and/or operations specified in the flowcharts anddiagrams may be re-ordered, processed serially, and/or processedconcurrently consistent with one or more embodiments. Moreover, any ofthe flowcharts and/or diagrams may include more or fewer nodes or blocksthan those illustrated consistent with one or more embodiments.

While all of the invention has been illustrated by a description ofvarious embodiments and while these embodiments have been described inconsiderable detail, it is not the intention of the applicant torestrict or in any way limit the scope of the appended claims to suchdetail. Additional advantages and modifications will readily appear tothose skilled in the art. The invention in its broader aspects istherefore not limited to the specific details, representative apparatusand method, and illustrative examples shown and described. Accordingly,departures may be made from such details without departing from thespirit or scope of the general inventive concept.

What is claimed is:
 1. A boost converter comprising: a clock generatorconfigured to output a clock signal; a switching converter configured tooperate at a frequency based on the clock signal; a hystereticcontroller configured to regulate an intermediate output from theswitching converter, wherein the hysteretic controller further includesa charge time to voltage converter configured to change the frequencycontrol signal opposite a change in charge time until a minimum chargetime of a capacitor detected; a power tracking module configured tochange a frequency control signal that is sent to the clock generator,the change in frequency is based on a current flowing into an outputcapacitor such that a charge time of the capacitor is minimized when thecurrent is maximized.
 2. The boost converter of claim 1 wherein theswitching converter is a switched capacitor converter.
 3. The boostconverter of claim 1 wherein the charge time to voltage converterincludes a first and a second capacitor, the first and a secondcapacitor are alternately charged such that as the first capacitordelivers charge to a load, the second capacitor buffers a boostedoutput.
 4. The boost converter of claim 3 further comprisingasynchronous hill climbing logic, perturb and observe logic, and anup/down counter.
 5. A method of adjusting a boost converter clockcomprising: by a controller, initializing a capacitor charge timevariable; decreasing a switching frequency of the boost converter; inresponse to a current charge time being less than the capacitor chargetime variable, update the capacitor charge time variable to keepdecreasing the switching frequency of the boost converter until currentcharge time exceeds charge time variable; increasing the switchingfrequency of the boost converter; and in response to a current chargetime being less than the capacitor charge time variable, update thecapacitor charge time variable to keep increasing the switchingfrequency of the boost converter until current charge time exceeds thecapacitor charge time variable; and changing, via utilizing a chargetime to voltage converter, a frequency control signal opposite a changein current charge time until a minimum charge time is detected.
 6. Themethod of claim 5, wherein the method includes the step of alternatelycharging a first and a second capacitor such that as the first capacitordelivers charge to a load, and the second capacitor buffers a boostedoutput.
 7. The method of claim 5, wherein the method includes changingthe frequency control signal in response to the capacitor charging time.8. The method of claim 7, wherein the change in the frequency controlsignal is opposite a change in capacitor charge time until a minimumcharge time of the capacitor detected.
 9. An apparatus comprising: aclock generator configured to output a clock signal; a switchingconverter configured to operate at a frequency based on the clocksignal; a hysteretic controller configured to regulate an intermediateoutput from the switching converter; and a power tracking moduleconfigured to change a frequency control signal sent to the clockgenerator; and a charge time to voltage converter that is configured tochange the frequency control signal opposite a change in charge timeuntil a minimum charge time of the capacitor detected.
 10. The apparatusof claim 9, wherein the hysteretic controller includes a boost converterthat includes a first capacitor and a second capacitor configured tobuffer the boost converter outputs in opposite phases.
 11. The apparatusof claim 9, wherein the hysteretic controller includes a boost converterthat includes a voltage regulator configured to utilize a modifiedDickson topology.
 12. The apparatus of claim 9, wherein the hystereticcontroller includes a boost converter, and the power tracking module isan asynchronous maximum output power tracking circuit that includes acapacitor bank, a Schmidt trigger comparator, and hill climbing logicconfigured to control the charging and resetting of capacitors in thecapacitor bank.
 13. The apparatus of claim 12, wherein the asynchronouscircuit is configured to switch when one charge period ends.
 14. Theapparatus of claim 9, wherein the change in frequency is based on acurrent flowing into an output capacitor such that a charge time of thecapacitor is minimized when the current is maximized.
 15. The apparatusof claim 9, wherein the apparatus includes a transducer modelled as aThevenin voltage source with an equivalent impedance, wherein thetransducer is connected to the switching converter.
 16. The apparatus ofclaim 9, wherein the hysteretic controller is a ping-pong hystereticcontroller.
 17. The apparatus of claim 9, wherein the clock generatorincludes a state machine and one or more multiplexers.
 18. The apparatusof claim 9, wherein the power tracking module is a maximum output powertracking module.